Hoffman Amplifiers Tube Amplifier Forum
Amp Stuff => Tube Amp Building - Tweaks - Repairs => Topic started by: Joe P on July 12, 2019, 06:38:57 am
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Hi,
Posted my first amp design a while back, https://el34world.com/Forum/index.php?topic=24808.0, and decided to follow the many advices to start simpler - so here's what I've come up with. Two-tube amp, single ended, EL84 output with a cascode frontend, James tonestack in the middle.
If my calculations are correct, plate voltage on the 12AU7 is too low to come close to its full gain potential, but I could sub a 12AY7 for a lot more gain if needed, i'd just change Ra to 47k.
Before anyone in the US wonders about the fuses on both hot and neutral, we have symmetrical plugs where I'm at.
Thoughts?
Edit: bypass and coupling caps are a bit large because I want to see if this is could be useable for bass as well.
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close to its full gain potential
never used the cascade setup, the 84 drives easy, with your 330 cathode R, I suspect you'll only need ~~ 5-7Vac at G1 to drive it to compression, maybe clipping.
G2 could probably be dropped to 470 - 1k, but the 1.5k won't hurt anything for getting it up and running.
enjoy the process, go slow, focus on clean, you'll have plenty of time to jam after :icon_biggrin:
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Lack of substantive response tends to indicate that you're still in the deep end of the pool, with no lifeguards on duty. My advice continues to be: build a proven design (cascode OK); produce a working amp; then mod it systematically.
Relatively few people have cascode experience.
Here's an except from Merlin: http://www.valvewizard.co.uk/cascode.html (http://www.valvewizard.co.uk/cascode.html)
N.B.: Your supply voltage is low. Note that for a typical gain stage, low supply voltage tends toward dark tone. Baxandall is dark tone (no mid dip).
What are your tube curves? What is your cascode's output impedance? What is the input impedance of the tonestack? How do the two balance?
What is the signal voltage into the EL84?
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Here on the forum, if I remember correctly, there was a guy that build a small amp with a cascode arrangement
may be someone can remember more than I and post a link to the thread
EDIT: I remembered the member (jbfumo) and think this is the thread
http://el34world.com/Forum/index.php?topic=14793.msg217649#msg217649 (http://el34world.com/Forum/index.php?topic=14793.msg217649#msg217649)
(http://el34world.com/Forum/index.php?action=dlattach;topic=14793.0;attach=59936;image)
Franco
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I'll agree with JJ here on B+ somewhat low. The 84 - modern, will do fine at 300 plate, so aim for 320 at 1st tap since you're gonna loose some to the cathode and G2.
Also agree on a deep read for the cascode, pretty sure on a conventional configuration, 1/2 tube before TS, 1/2 tube after you can hit a target of ~~ 7Vac to the 84's G1.
I personally like to take my SE PA tube B+ from the 2nd PS tap using FWB and ~ 100uf 470ohm pi filter before feeding the PA tube. There is no inherent hum cancelling like you get in PP design
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Your PT is rated for 190V at 75mA and your rig looks like it isn't going to use much more than 49mA, so I am going to use a conservative value of 260V for your B+. 7mA of screen current and 5mA of cascode current both go through the 470r dropping resistor. By Ohm's law, 12mA times 470 Ohms is 5.64V, so the second node will be about 6 Volts less than the first node. Only the 5mA of cascode current goes through the 2K2 dropping resistor for a voltage drop of 11 Volts, so the third node will be about 243V.
The gain of the cascode is about the same at most supply voltages, so 200V (or 243V) will allow it to come close to its full gain potential. Headroom will decrease with decreasing supply voltage, but this cascode is the first stage so that headroom is not a consideration. A 12AY7 cascode with the same components as a 12AU7 cascode is likely to yield similar gain.
As far as bass is concerned, C5 needs to be changed to 22uF and C7 needs to be changed to 100nF. The existing 1uF/560r cathode combination has a cutoff frequency of 284Hz, so there is significant bass rolloff. C7 at 10nF will also contribute a little to bass rolloff.
The gain of the cascode with a 10K plate resistor is probably around 18. If you put a 100mVp guitar signal into it, you will only get 1.8Vp out of it and that is at higher frequencies. Even without the tone-stack loss, this isn't enough to drive the EL84. Changing the plate resistor to 47K will probably give you a gain of 60 for an output of 6Vp, still too small after the tone-stack loss. A 100K plate resistor might give you, who knows, a gain of 90? Still might not be enough to drive the EL84 after tone-stack loss.
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Hi,
Posted my first amp design a while back, https://el34world.com/Forum/index.php?topic=24808.0, and decided to follow the many advices to start simpler - so here's what I've come up with.
And it is simpler! That's good, but I'm going to suggest that you also think flexibler.
I'll spell it out: Buy a chassis and power supply for a Marshall 1974 (a.k.a. "18 Watt".) Both are overkill for an SE EL84 build, but they'll provide room to grow. Here in the US we might find an amp from an old Hammond organ for cheap, or even a whole organ for free. I don't know how common organs are where you live, but you might be able to find something. Or just buy the 18 Watt chassis and transformer.
Then build this, more or less exactly as drawn (except for the rectifier.): http://mercurymagnetics.com/images/pdf/schematics/wiring/E-VJ-schem1a.pdf (http://mercurymagnetics.com/images/pdf/schematics/wiring/E-VJ-schem1a.pdf)
It should take very little time, but I guarantee that if this is your first build, then you will learn a ton. But, It'll be a lot easier troubleshoot because you'll know what to expect from the circuit. Or, if you don't then everyone here will and it'll be easy to get help.
Once you have that working, then go nuts trying different tubes, cascodes, whatever... There're enough tube sockets that you can have multiple preamps at once. Eventually you can try PP if you want. You'll just need to find another output transformer. All with that chassis and power transformer.
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> If my calculations are correct, plate voltage on the 12AU7 is too low to come close to its full gain potential,
How are you calculating??
SPICE figures your plan has gain of about 6 (yes, six!) and eats near 5mA current.
If 10k is changed to 100k gain rises to 11 and current falls to 1.5mA.
A single section of 12AU7 can give gain of 16. Two in cascAde gives gain over 200. At total current near 3mA. And much lower impedances.
The usual rule-of-thumb for a guitar amp is two high-Mu stages cascade; maybe more if there is a tone-stack.
Yes, cascOde 12AU7 has Mu like 20*20=400, but (like pentodes) you can not get a load impedance high enough to use all this. (Even so, 6 seems absurd.)
As a general rule, a Power Tube needs a fairly direct feed from a driver. You drive through the quite lossy James tonestack. Even with a super-hot guitar to cover the low gain, it is likely the pre-tone stage will distort before the power stage makes its big 4 Watts.
Gain does not vary a lot with plate voltage, only slowly with supply voltage. Overload does vary pretty directly with these voltages of course.
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Some more cascode stage examples attached FWIW. (The one on the R.H. side in the Weber schematic is not a cascode, but is a totem pole SRPP with the upper triode acting as a current source in place of a load resistance, so another interesting variant on totem pole designs)
These would all benefit from moderate (30V - 40V) heater elevation, especially the AU7 examples
Another good triode to use for a cascode would be a 12WD7, with the AU triode as the input triode, and the AX7 as the grounded-grid (upper) triode. This should enable good gm for the input triode and reasonable output signal swing on the output triode.
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12WD7
no such tube. 12DW7/7247 - is that the tube you're thinking of?
--pete
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12WD7
no such tube. 12DW7/7247 - is that the tube you're thinking of?
--pete
WD40 = engine oil and I've just been putting some in the car before I posted this duh.
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Thanks for the replies!
It's evident that my calculations on that cascode are completely wrong. It's Merlin's guide I've been using. I'm going to figure out where I went wrong, and then go with Tony's suggestion to start with a Valve jr and a 1974 psu.
No cheap Hammonds here, but I have a couple old transistor amps that I can butcher.
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> calculations on that cascode are completely wrong. It's Merlin's guide I've been using. I'm going to figure out where I went wrong
One error found. Upper grid-cap is actually a cathode cap. I though that was odd, but didn't get-it until I put your plan against Merlin's.
With that grid just following cathode this is not a cascode but an idle resistor in the plate circuit.
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With the correct grid connection, gain rises from ~~6 to like 18. Your undersized capacitors (all around) give a major bass-loss.
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With fully ample caps and 47k load, gain soars to near 60 and full bass. (You may want less bass for guitar, but let's see full response before we start carving.)
This uses a whole bottle to do what a half a bottle of 12AX7 would do, and about twice the current of a single 12AX7 stage.
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One error found. Upper grid-cap is actually a cathode cap. I though that was odd, but didn't get-it until I put your plan against Merlin's.
Ahhh, I totally missed that. I was wondering how you got a gain of 6 while I got a gain of 18.
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The ultimate voltage gain of a cascode should be Mu*Mu. We never get there (in audio) because of loading and DC-feed impedances.
Curious, I tried two way-out extremes. A 1,000H choke (you can't buy one cheap or full audio band); and a huge resistor with a huger supply voltage, both with impossibly large loads.
The big choke with 250Meg load does come very close to Mu*Mu. And it promises a voltage swing like 140V. Which makes the input overload near 140V/400= 0.35V, marginal for e-guitar. Of course the choke is nearly unobtainium, and the "250Meg" AC load needs buffering. Since two-half 12AX7 makes gain over 2,000 and can drive 200k comfortably, why?
Resistor loaded 1Meg with 1KV supply doesn't get halfway there. According to G.E., Gm of 12AU7 at 100V 1mA is like 600uMho, or 1/1700 Ohms. Mu here has fallen to 13. So "plate resistance" of the cascode is near 280k, and "Mu" is 169. Which does not quite match what the Koren model tells me. But IAC the phantom of high gain is clouded by uselessly high output impedance.
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> calculations on that cascode are completely wrong. It's Merlin's guide I've been using. I'm going to figure out where I went wrong
One error found. Upper grid-cap is actually a cathode cap. I though that was odd, but didn't get-it until I put your plan against Merlin's.
With that grid just following cathode this is not a cascode but an idle resistor in the plate circuit.
Ah, yeah, and it's the correct circuit that I've made calculations on.
With the correct grid connection, gain rises from ~~6 to like 18. Your undersized capacitors (all around) give a major bass-loss.
I must have made a calculator typo on the cathode cap. For Ck on the EL84, if that's too low my definition of "roll off" can't be the same as Merlin's.
Ahhh, I totally missed that. I was wondering how you got a gain of 6 while I got a gain of 18.
I got 26, so there's something else I've been doing wrong.
Getting as much gain as possible wasn't really my goal here, I wanted the pentode sound without the microphony. I've only heard pentode preamps, not played one, so I'm curious.
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as much gain as possible wasn't really my goal here
agreed, what you need is enough to overcome the circuit n TS loss (I don't do math:), and have enough to at least bend the 84. Most don't really care for 84's that are screaming, but bent is goooood :icon_biggrin:
from what 2Deaf came up with and my ballpark, I'd aim for 8VACp-p on G1
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The 12DW7 variant I mentioned earlier - should be good for driving a TMB tone stack as well as having a decent amount of gain. To make it gnarlier, change to grid-leak bias (using 1M) on the AX triode (or experiment with reducing the upper triode grid bypass cap size, or increasing the load resistor (slightly) for the upper triode).
Wait til you get it installed, then decide whether it needs heater elevation or not. It shouldn't need any but you could always give it a little boost by running the heater 'ground reference' to the cathode of your cathode biased output tube.
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I re-did my ballpark with #'s from datasheet.
If their correct, you might need more drive than ~~8vac.
I use the cathode volts DC as a guide for Vac on G1. so closer to 20vac to get the tube bent hard
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The 12DW7 variant I mentioned earlier - should be good for driving a TMB tone stack as well as having a decent amount of gain.
I can't see how you are going to get much more than unity gain out of that rig. Also, I'm not catching on to the advantage of using a 12AX7 for the upper triode.
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...upper triode grid bypass cap...
And yet, again, the cap is connected to cathode not grid. The upper tube is an inert resistor.
I also too have doubt that a high-Mu upper tube really wrings any more gain out of a cascode.
Moreover for same current, a 12AX7 has more Gm than a 12AU7.
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The 12DW7 variant I mentioned earlier - should be good for driving a TMB tone stack as well as having a decent amount of gain. To make it gnarlier, change to grid-leak bias (using 1M) on the AX triode (or experiment with reducing the upper triode grid bypass cap size, or increasing the load resistor (slightly) for the upper triode).
Wait til you get it installed, then decide whether it needs heater elevation or not. It shouldn't need any but you could always give it a little boost by running the heater 'ground reference' to the cathode of your cathode biased output tube.
cap connects as shown in attached.
--pete
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How about a regular gain stage on the input and a cathode follower after the cascode to drive the TS?
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cap connects as shown in attached.
That certainly makes a lot more sense, but why 470nF? Looks like it will take a long time to charge when B+ is first applied during which time the gain will be reduced due to the lower "screen" voltage. It seems to me that 10nF would be plenty for a flat response.
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How about a regular gain stage
wait, the numbers are still coming in :laugh:
the bottom-line, determining enough signal to drive the PA, and whatever "tone" flavor you want to add before that.
Me, I'd stick with your idea but expect, work out, a 2nd pre tube in your design, just in case you need more gain or impedance matching (CF)
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FWIW: https://sites.google.com/site/stringsandfrets/Home/deluxe-plus (https://sites.google.com/site/stringsandfrets/Home/deluxe-plus)
Contains an interesting article by Aiken.
Note that the cascode drives a cathode follower > tonestack. Followed by a cascode tone recovery stage driving another cathode follower!
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but we all agreed to KISS :icon_biggrin:
I do think a 2nd tube might be needed, a 3rd for an SE EL84 is overkill by alot
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KISS!?! I thought the design might turn Joe P ON. But seriously, folks, if you want to KISS, the design suggests the desirability of a CF after the cascode to deal with the cascode's hi output impedance; along with the "need" for tone recovery stage. (Not to mention Aiken's insights).
Also, if Joe P wants an example of a proven, complex cascode design, there it is, with soundtracks. (Not that I recommend it as a first tube build.)
Not to mention that if a cascode design complies with KISS, then we must already be on our 3rd date. Someday, we may even give birth to an actual design.
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cap connects as shown in attached.
That certainly makes a lot more sense, but why 470nF? Looks like it will take a long time to charge when B+ is first applied during which time the gain will be reduced due to the lower "screen" voltage. It seems to me that 10nF would be plenty for a flat response.
i didn't choose that value - 47nF could be overkill and is 10n enough though? personally i'd use 100nF just because john broskie does. :p
--pete
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i didn't choose that value - 47nF could be overkill and is 10n enough though? personally i'd use 100nF just because john broskie does. :p
I'm sorry, I knew you didn't choose 470nF and I wrote it wrong so that it implied that you did.
With the usual equation for cut-off frequency, 1M and 10nF would yield 16Hz. But I'm not well enough versed on tube cascodes to state that the upper grid leak resistor and bypass cap will act like that. The charge/discharge path for the 1M grid leak makes me suspect that it will appear to be larger than 1M for cut-off frequency purposes. Really going out on a limb with that one.
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The cap that clamps the "idle" grid in a LTP or a Cascode has to be larger than you think.
A coupling cap we hardly notice 29% loss (-3dB).
These "idle" grids have to be IDLE. Even a few-percent wobble degrades expected performance.
You often see, on LTPs, 0.01u on one grid and 0.1u on the other. This 10X difference is a good rule of thumb.
Somewhere in this thread (probably a link) someone used 1uFd. Yes, a 1 Second time constant just to get to 67% of final value. It isn't far off from the 10k+40uFd (0.4sec) found in B+ decoupling networks. This is probably moot in light of 11 seconds for heater warm-up and 234 seconds to adjust your shorts, find your pick, adjust your strap......
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I would not want to troubleshoot that DR...
But as you say, it is a proven cascode circuit. Now, I am going to KISS at first, but it feels good to have a plan for the future!
My thinking is, first get the Valve jr/1974 up and running, then mess around with tonestack and other tweaks, and then maybe insert the cascode.
Regarding the double cascode/CF, he's driving a PP pair of 6V6s with NFB, I'm only driving a SE EL84 - so my remaining half 12AX7 could be more than enough, right? Question is whether to have it as a recovery stage or as an input stage to overdrive the cascode.
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fwiw
I use this circuit as a "standard"
http://www.valvewizard.co.uk/dccf.html
with small bottle PA I like to put it pre TS, ala Marshall, with big bottle I use it after TS as recovery/drive for PA
pre gets to crunch nicely, less "annoying" pre-amp distortion, post keeps the signal "original" enough with the extra demands of a large PA tube
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I would not want to troubleshoot that DR...
But as you say, it is a proven cascode circuit. Now, I am going to KISS at first, but it feels good to have a plan for the future!
My thinking is, first get the Valve jr/1974 up and running, then mess around with tonestack and other tweaks, and then maybe insert the cascode.
Regarding the double cascode/CF, he's driving a PP pair of 6V6s with NFB, I'm only driving a SE EL84 - so my remaining half 12AX7 could be more than enough, right? Question is whether to have it as a recovery stage or as an input stage to overdrive the cascode.
The beauty of that amp is not to miss the trees for the forest! I.e., the over-complicated forest of that amp contains some really useful, simple clusters of trees. These can be effectively plagiarized.
1. A working Cascode-CF input section.
2. The CF drives, in essence, a Fender TMB tonestack. (Like the old Bassman circuit)
3. A working Cascode-CF tone recovery stage.
So, there's the option to use just one, or both, cascode circuits.
From the tube charts: At 360 Plate volts a pair of 6V6s in PP needs -22.5 fixed bias volts, or +22.5 cathode bias volts. @ 250 Plate volts, an SE EL84 needs 7.5V bias. You can plagiarize the cathode bias voltage from a known SE EL84 amp @ your plate voltage; and feed signal voltage accordingly.
See, e.g., Epiphone Valve Jr., which I think was suggested earlier, as the basis for your project.
This has Dummyload's breadboard written all over it!
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See, e.g., Epiphone Valve Jr., which I think was suggested earlier, as the basis for your project.
Yup, that's my starting point - though I've started contemplating maybe just using the PA and taking the pre from some version of Champ. Nicer sound to start with...
This has Dummyload's breadboard written all over it!
Can't find it with a quick search... I've been planning on whipping something up with screw terminals, I'll be sure to dig through the threads I did find for further layout ideas
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another approach - in theory it should work... maybe split the second cascode into cascaded stages? please comment liberally as i am new to cascodes as well. i used joe_p's presented plan as the basis for this design, except i used higher B+ and EL34 because i don't like EL84 amp and ~10W of power for some added clean headroom.
just my opinion, but 12AU7s aren't of much use in guitar amps (flame away h8rs)*, except for use in high power amps that require low Z drivers, and the odd-ball amps that use it for a reverb driver, etc.. they are rarely used as the main gain stages of a guitar amp. could the cascode change things? maybe.
i have not built, nor have i breadboard the circuit presented, just throwing out other options for those that may want to experiment with low mu tubes and the tonal characteristics that are a result of cascode gain stage(s).
regards,
--pete
if you wish to add NFB, then add a 100R under R12 and C9 then connect a 10K from the 8 ohm tap to the junction of R12 C9 and the 100R - the 10K is a value that you would use your own ears to choose. 1K to 10K is a suitable range to start with. use the princeton reverb schematic for NFB insertion as a reference if the preceding is unclear.
* if i'm looking to use the 12AU7 in any application, i almost always turn to 6CG7 or the 12BH7. IMO, a much better tubes + i have a box full.
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just my opinion, but 12AU7s aren't of much use in guitar amps
My main reason for using the 12AU7 here was that that's what Merlin suggests for use in a cascode - now I'm thinking I'm just going to go with a 6922 for simplicity - I can just copy the cascode stages from that DR from hell and tweak from there.
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h8ter alert :icon_biggrin:
i have a box full.
my 1st reason for using AU7
my 2nd, I am blissfully ignorant of some tube math, so I just plug it in an observe. What I find for "V1", because it's a low gainer, signal only gets a smallish boost, but way more than the noise floor, so it leaves more hissey/hummy out of the signal path.
In a DCCF it seems to be harder to slap around, thus adding less sour taste to the signal
I wouldn't try and use those arguments as an answer to your EE tube Final exam :icon_biggrin:
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I had some time today so I built a cascode stage and ran some frequency response trials with various capacitors in the Cg2 and CK positions. Attached are the test setups that I used.
I set Ck at 220uF for the Cg2 tests so that the cathode would be fully bypassed and not a factor in the frequency response. I found that the cut-off frequencies at the output closely followed 1/(2*pi*R*C).
I set Cg2 to 0.22uF for the Ck tests so that the "screen" had a flat response. I found that the cathode bypass capacitor has a much more potent effect in a cascode than in a standard gain stage -- like twice as much. So the cut-off frequencies at the output more closely follow 1/(pi*R*C).
That's what I got, anyways. Maybe somebody can see if they can replicate my results.
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I had some time today so I built a cascode stage and ran some frequency response trials with various capacitors in the Cg2 and CK positions. Attached are the test setups that I used.
I set Ck at 220uF for the Cg2 tests so that the cathode would be fully bypassed and not a factor in the frequency response. I found that the cut-off frequencies at the output closely followed 1/(2*pi*R*C).
I set Cg2 to 0.22uF for the Ck tests so that the "screen" had a flat response. I found that the cathode bypass capacitor has a much more potent effect in a cascode than in a standard gain stage -- like twice as much. So the cut-off frequencies at the output more closely follow 1/(pi*R*C).
That's what I got, anyways. Maybe somebody can see if they can replicate my results.
Very interesting! Merlin Blencowe (where I've gotten much of my information) says: For most purposes it is sufficient to choose the cathode bypass capacitor according to:
Ck = 1/ (2 . pi . f . Rk)
So now we have empirical evidence that says otherwise.
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> I found that the cathode bypass capacitor has a much more potent effect in a cascode than in a standard gain stage -- like twice as much. So the cut-off frequencies at the output more closely follow 1/(pi*R*C).
The cathode node impedance is not just the 1K resistor, but also the internal cathode impedance. Which is frequently very similar to the bias resistance. So not 1k but 500r. Which is about your lost factor of 2.
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Very interesting! Merlin Blencowe (where I've gotten much of my information) says: For most purposes it is sufficient to choose the cathode bypass capacitor according to:
Ck = 1/ (2 . pi . f . Rk)
So now we have empirical evidence that says otherwise.
Merlin never said that his equation is for a frequency that gives a gain that is 3dB below fully-bypassed gain, but mine is.
In general, if you come up with something that doesn't agree with what Merlin said, then you are either in error or you misinterpreted what he said.
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Merlin Blencowe discusses cascode design in a couple of his books (as well as small signal pentodes).
The extent to which the screen (in pentodes) and the upper triode grid (in cascodes) is bypassed, has a significant impact on bandwidth. If the upper triode grid is ‘fully’ bypassed, the lower triode acts purely as a transconductance triode - because the effect of ‘fixing’ the upper triode’s grid voltage (at ‘all’ frequencies) is analogous to the way operation of a screen (in pentodes) shields the input (signal) grid from changes in output voltage at the plate. Therefore, bypassing the lower triode’s cathode resistor does not have as much of an effect on bandwidth, as fully bypassing the upper triode’s grid does. The reason I used 470nF in my earlier example, is because this works better than 100nF in this regard. (However, if you want maximum gain, then you should bypass the cathode resistor for the lower triode as well).
If you want maximum current, then use grid leak biasing on the upper triode.
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The cathode node impedance is not just the 1K resistor, but also the internal cathode impedance. Which is frequently very similar to the bias resistance. So not 1k but 500r. Which is about your lost factor of 2.
There is no lost factor of 2, but there is a lost interpretation of the data.
1/(pi*R*C) only works for the Test Setup with a 1K cathode resistor. If you change the cathode resistor to 2K, then 1/(0.6*pi*R*C) is a better estimate for the cut-off frequency. If you change the cathode resistor to 560, then 1/(2*pi*R*C) is a pretty good estimate.
For a partially bypassed cathode, the frequency at 1/(2*pi*R*C) is where the gain has risen a certain amount above the unbypassed gain. If you choose 3dB for that certain amount above minimum gain and there is a 6dB difference between unbypassed and fully bypassed gain, then the -3dB cut-off frequency is the same as the +3dB above minimum gain frequency. With a 560r cathode resistor in the Test Setup, the gain spread is right in there at 6dB and that is why 1/(2*pi*R*C) works for the -3dB cut-off frequency.
If you change the cathode resistor to 2K, the difference in gain between unbypassed and fully bypassed is more like 10dB. Now the +3dB above minimum gain frequency is not the same as the -3dB cut-off frequency. There is a 6dB/octave section in the S-Curve plot for the frequency response between the two 3dB points. So the -3dB cut-off frequency occurs at a higher frequency than the +3dB above minimum gain frequency. The denominator in the cut-off frequency equation needs to be reduced to account for the difference in 3dB points.
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Therefore, bypassing the lower triode’s cathode resistor does not have as much of an effect on bandwidth, as fully bypassing the upper triode’s grid does.
Can you re-word that? I can't make any sense out of it as it stands.
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another approach - in theory it should work... maybe split the second cascode into cascaded stages? please comment liberally . . .
The 1.2K/10uF cathode combination looks like it might have a cut-off frequency around 13Hz, but it probably has around a 30Hz cut-off frequency with this cascode. You may have already known that because 30Hz sounds pretty good to me for a guitar amp.
The Miller effect has been nearly eliminated because the bottom triode has very little gain (this is about the only reason I can see to use a cascode with tube triodes). If I generously give the bottom triode a gain of 1.5, then the input impedance will be 1.5pF times the 1.5 gain plus the grid-to-cathode capacitance of 1.5pF. Call it 4pF. So an 820K grid stopper will give about the same high-frequency cut-off as 33K with a standard 12AX7 gain stage.
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If you change the cathode resistor to 2K, then 1/(0.6*pi*R*C)
I can't make any sense out of it as it stands.
not looking for a deep dive, simple bellyflop works :icon_biggrin:
the 2K, isn't that the "R" in the equation?
If so, why .6
If not the R, why
thanks
dave
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the 2K, isn't that the "R" in the equation?
If so, why .6
If not the R, why
Yes, 2K is the "R" in the cut-off frequency equation. 0.6 comes from observed frequencies when the the output voltage was down 3dB from the maximum voltage. I just re-arranged the original equation to solve for the factor that R*C had to be multiplied by to get those frequencies. Then I wrote that factor as 0.6*pi instead of just 1.9 for dramatic effect. For example: I observed a -3dB voltage at 26Hz with a 2K/10uF cathode combination. The factor I am looking for is 1/(26Hz*2K*10uF) which is 1.9 and that converts to 0.6*pi.
You get 8Hz if you plug the same numbers into 1/(2*pi*R*C). This is not the -3dB cut-off frequency, but rather the point when the gain is up somewhat from the un-bypassed gain. The frequency difference between the two points depends on the value of the cathode resistor, all other things being equal.
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Therefore, bypassing the lower triode’s cathode resistor does not have as much of an effect on bandwidth, as fully bypassing the upper triode’s grid does.
Can you re-word that? I can't make any sense out of it as it stands.
If you were to bypass either only the cathode of the input triode, or the grid of the output triode, then the latter would have more effect on gain and bandwidth.
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If you were to bypass either only the cathode of the input triode, or the grid of the output triode, then the latter would have more effect on gain and bandwidth.
Thanks.
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from observed
that was my working hypothesis :laugh:
so, math says in ideal 1/2pirc, but the real world's never ideal :icon_biggrin:
thanks
dave
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Shooter, that's the wrong conclusion. For a standard passive RC-filter, cutoff frequency, the frequency f for which the gain is 3dB lower than the gain in the pass band, is: f = 1/ (2*pi*r*c). This is carved in stone.
But a tube is a reactive, not a passive device. It does not behave like a fixed resistor, especially for AC operation. So, the passive formula does not fully apply to bypassing a tube element with a cap.
Added complexity: Merlin states that the performance of actual cascode circuits may not conform to design modeling.
So 2deaf is building & measuring to find out what's actually happening.
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This is carved in stone.
got the tablets :laugh:
you're still calculating a passive RC with Rk/Ck, you just need more math to calculate tube dynamics, then put the answers in a yatzee shaker n pour out :icon_biggrin:
OR
as 2Deaf shows, n my preferred method, observe using math to insure you're watching the correct channel :laugh:
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cap connects as shown in attached.
That certainly makes a lot more sense, but why 470nF? Looks like it will take a long time to charge when B+ is first applied during which time the gain will be reduced due to the lower "screen" voltage. It seems to me that 10nF would be plenty for a flat response.
I missed this question before.
Longer charging time also equals longer discharging time, which equals shunting lower frequencies.
Also, everything in the amp takes a while to charge up at startup, so there is nothing unusual about that
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Also, everything in the amp takes a while to charge up at startup, so there is nothing unusual about that
Not to mention the 234 seconds to adjust your shorts and such.
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Yeah,so...I looked at equation (2) in Chapter 12, Section 2 of RDH4 for a minute, maybe a minute and a half. My motivation crisis has become so severe that I'm not even going to try setting A'/A to 0.707 and rearranging the equation to solve for frequency. So I decided to run with what PRR said in #42. Seems to work for triode gain stages, but what about cascodes? I re-worked the cut-off frequency equation a little.
fc' = 1/(2*pi*Rk' *C) Where fc' is the -3dB cut-off frequency and Rk' is (Rk x rk) / (Rk + rk). A slight re-arrangement of that equation gave me:
fc' = (Rk + rk) / (2*pi*Rk*rk*C)
I determined the internal cathode resistance for the different operating conditions and plugged in the numbers. Gives a real good fc' for the 1K and 2K cathode resistors and close-enough results with the 560r cathode resistor.
(edit: I have apparently angered the font size gods)
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I like it font fail n all :laugh:
despite my mathobia I enjoy seeing it played out.
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I started looking at Merlin's Fixed Biasing the Upper Triode on Valve Wizard's Cascode pages. I just can't see how Cg2 is going to have any effect on the frequency response when there is no audio current in the fixed bias circuit. What am I missing?
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I started looking at Merlin's Fixed Biasing the Upper Triode on Valve Wizard's Cascode pages. I just can't see how Cg2 is going to have any effect on the frequency response when there is no audio current in the fixed bias circuit.
Do you mean this page, image, text?
http://www.valvewizard.co.uk/cascode.html
(http://www.valvewizard.co.uk/Cascode2.jpg)
"If we choose a value of 560k for R1 and .... R2 ....= 104k ....85k.... This value is used to find a suitable decoupling capacitor. So, for a low roll-off of 20Hz...
I think his fingers got ahead of his mind. The grid is so high impedance that 85k is "zero" out to 20kHz (assume 100pFd Miller C).
But this point "should" be bypassed or it will suck-up crap from adjacent parts and the room; also power supply crap. The "required" value depends on how much crap is around and what your final-test specs are. My gut says it could maybe be less than 0.1uFd, but that's a common part and a smaller value would not save dollars. If you build by the millions, pennies count, better think carefully. But in DIY, 0.1uFd seems a fine value, even if the explanation there is misleading.
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Do you mean this page, image, text?
http://www.valvewizard.co.uk/cascode.html
(http://www.valvewizard.co.uk/Cascode2.jpg)
"If we choose a value of 560k for R1 and .... R2 ....= 104k ....85k.... This value is used to find a suitable decoupling capacitor. So, for a low roll-off of 20Hz...
Yes, that's the one.
0.1uFd seems a fine value, even if the explanation there is misleading.
I figured I had misinterpreted it. Once again, thanks.
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Cg2 anchors the g2 voltage under signal conditions (according to the frequency point that the capacitance of the cap can usefully ‘fix’). Without Cg2, the grid would fluctuate slightly, as current through the upper triode changes, because the shunting function of the g2 voltage divider isn’t perfect enough to override the effect of changes in g2 grid current. It is analogous to bypassing (or decoupling) the screen supply in a pentode.
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Cg2 anchors the g2 voltage under signal conditions (according to the frequency point that the capacitance of the cap can usefully ‘fix’). Without Cg2, the grid would fluctuate slightly, as current through the upper triode changes, because the shunting function of the g2 voltage divider isn’t perfect enough to override the effect of changes in g2 grid current.
Merlin says, "The upper grid does not draw any current, so we require only a voltage reference . . . " This has been found to be true and Cg2 has no effect whatsoever right up to the point of grid-current clipping. There are actually two grid-current clipping events and Cg2 affects the onset point of both of them. It only has a subtle effect on the first onset point with a much larger effect on the second onset point.
It is analogous to bypassing (or decoupling) the screen supply in a pentode.
Not really. We fully bypass the pentode screen in order to keep gain constant and we fully bypass the cascode "screen" in order to prevent unwanted noise at g2.
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um. I'm the pointy hat kid in back;
We fully bypass the pentode screen in order to keep gain constant and we fully bypass the cascode "screen" in order to prevent unwanted noise at g2.
so far I'm holding on, but the "unwanted" crap/noise at the fake g2 - does that hold true in a real pentode where the cap "keeps gain constant", is it also keeping crap out as a bonus?? :dontknow:
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... but the "unwanted" crap/noise at the fake g2 - does that hold true in a real pentode where the cap "keeps gain constant", is it also keeping crap out as a bonus??
I don't actually buy into the noise at g2 thing, I'm just playing along. The gain for a signal injected at g2 of the cascode is very low as compared to the gain for a signal injected at g1. The same is true for a pentode where the gain using the screen grid is way lower than the gain using the control grid. The lack of amplification for the noise when injected at the "screen" in either case really diminishes the concern about the noise. But, yeah, I'm sure the real pentode is reaping a bonus by having the bypass capacitor lowering the crap level.
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It is analogous to bypassing (or decoupling) the screen supply in a pentode.
Not really. We fully bypass the pentode screen in order to keep gain constant and we fully bypass the cascode "screen" in order to prevent unwanted noise at g2.
Well yes, ... noise that would otherwise be caused by tiny fluctuations in grid current. If you hold g2 at a constant voltage, and raise and lower the cathode voltage, you will respectively raise and lower the tube current. Most of that current will be sourced through the plate. A smaller amount will be sourced through the mechanism that is used to fix Vg2. The voltage divider used for fixing the g2 bias is a crude shunt stabiliser, which ‘drowns out’ most, but not all, of the (few nano-amps of forward and reverse) g2 current. The cap stabilises Vg2 for the bit that isn’t drowned out, stopping noise.
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so, math says in ideal 1/2pirc, but the real world's never ideal :icon_biggrin:
There was a very smart man on the Aussie Guitar Gearheads forum, who had a signature line I really liked: "If theory and practice don't agree, you haven't used enough theory."
That simple 1/(2 pi R C) formula works perfectly for a single RC high pass or low pass filter. In the real world, if you use a 1% resistor and a 1% cap, and put them between two op-amp buffers, the real-world frequency response will be within a couple of percent of the theoretical 1/(2 pi RC) formula, over the entire frequency range which the opamps handle well.
But a single resistor and a single cap isn't what we have at the cathode of a triode gain stage. We don't get a simple high pass filter response. Instead, we have a shelving filter, which has *two* corner frequencies - flat at very low frequencies, flat at much higher frequencies, and with a sloping "ramp" connecting those two flat regions. This is because there is more than one resistance involved: not just the external cathode resistor, Rk, but also other invisible resistances inside the tube.
What matters isn't Rk, then, but rather (as PRR said earlier), it is the resistance "seen" at the cathode if you measured it. This is a combination of the external cathode resistor Rk, the internal cathode resistance (same as 1/gm, sometimes called rk), and in the case of a triode, also on external *anode* resistance Ra (aka Rp), the internal anode resistance ra (aka rp), and the mu of the tube!
With a 12AX7, for instance, if we take the nominal mu of 1600 microamps/volt, then the internal cathode resistance is (1/mu), or about 625 ohms. This is not ten times bigger than the typical 1500 ohm external cathode resistance, nor is it ten times smaller. That means we can't just ignore one resistor when calculating corner frequencies caused by the cathode bypass cap: we really have to include both resistances, and that means a considerably more complicated formula than just plain old 1/(2 pi R C).
These days it isn't hard to use a computer to simulate the frequency response, but if we're talking about guitar gain stages which have already been built thousands of times before, why bother? I only bother calculating if I'm using an oddball tube that nobody seems to have used for a guitar amp before, which might need rather different external resistors and cap values to produce the desired frequency response.
And it's really nice to have an actual frequency response measurement, if only to confirm that we have, in fact, used enough theory...so just recently I splurged some fifty bucks ($CAD) on a Syscomp CGM-101 ( https://www.syscompdesign.com/product/cgm-101/ ).
That must have been a closeout deal, as the CGM-101 seems to be gone for good. But it's pretty much exactly what I wanted - eleven bits of vertical resolution but a small 200 kHz bandwidth, useless for today's fast digital circuits, but pretty much what we want for audio measurements. 11 bits of vertical resolution is much better than the somewhat-affordable digital 'scopes like my Rigol, which only has a rather pitiful 8 bits (256 steps) of vertical resolution, but has far greater bandwidth (which is useless for audio.)
I cannot vouch for it's accuracy, but there is an online cathode bypass cap calculator here: https://www.ampbooks.com/mobile/amplifier-calculators/cathode-capacitor/calculator/
The attached image shows the result of running that calculator on a half-12AX7 with a 1uF cathode bypass cap, and the usual 1.5k Rk and 100k Ra (aka Rp). As you can see, it's neither a high pass nor a low pass filter, but instead, a shelving filter with two corner frequencies.
-Gnobuddy
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you just need more math
:icon_biggrin:
what's the 'ol lyric, different strokes for different folk
I like this formula;
(time/fun) * $ = theft of others + (individual tweaks)
-------------------------------------------------------------------------------------
musical happiness
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I like this formula;
(time/fun) * $ = theft of others + (individual tweaks)
-------------------------------------------------------------------------------------
musical happiness
Agree very much about the musical happiness. :smiley: Just this last Monday (a holiday here in British Columbia), I spent a day of musical happiness at an outdoor jam, held under big shady trees on a beautiful hobby farm in the city in which I live. I met several good musicians, got to make music with several of them, enjoyed the food (potluck) we'd all brought to share, played with a cute dog, and even got to pet a friendly foal who came right up to the fence to have his forehead scratched. It was a wonderful day!
Lately, musical happiness is the motivation for most of the math I'm actually bothered to do - it's almost always in the quest to understand guitar sound better!
-Gnobuddy
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:think1: :think1: :think1:
It should be the inverse of:
(time/fun) * $ = theft of others + (individual tweaks)
-------------------------------------------------------------------------------------
musical happiness
otherwise the more musical happiness the smaller the outcome :cussing:
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But a single resistor and a single cap isn't what we have at the cathode of a triode gain stage. We don't get a simple high pass filter response. Instead, we have a shelving filter, which has *two* corner frequencies - flat at very low frequencies, flat at much higher frequencies, and with a sloping "ramp" connecting those two flat regions. This is because there is more than one resistance involved: not just the external cathode resistor, Rk, but also other invisible resistances inside the tube.
A shelved response is not the same as a shelving filter. A high-pass filter has a shelved response above the cutoff frequency, but it is not a shelf filter. A low shelf filter also has a shelved response above the cutoff frequency, but what happens below the cutoff frequency isn't necessarily the same as a high-pass filter. A low shelf filter affects the frequencies below the cutoff frequency by either cutting them or boosting them. So the term "high-pass filter" tells you what is going to happen to the frequency response and the term "low shelf filter" only tells you where the frequency response is going to deviate from flat.
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With a 12AX7, for instance, if we take the nominal mu of 1600 microamps/volt, then the internal cathode resistance is (1/mu), or about 625 ohms. This is not ten times bigger than the typical 1500 ohm external cathode resistance, nor is it ten times smaller. That means we can't just ignore one resistor when calculating corner frequencies caused by the cathode bypass cap: we really have to include both resistances, and that means a considerably more complicated formula than just plain old 1/(2 pi R C).
mu is not 1600uA/V, transconductance (gm ) is. 1/gm is not the internal cathode resistance of a common-cathode triode gain stage, but it is an approximation of the internal cathode resistance for a cathode follower. The internal cathode resistance for the common-cathode triode gain stage is (Rp + rp ) / (u+1).
One time I calculated the parallel impedance of Rk and Ck for a bunch of frequencies and plugged that value into the gain formula in place of Rk . I converted the results to dB using the fully-bypassed gain as the reference point and graphed it. I then took an actual 12AX7 and observed the actual gain at those frequencies and graphed that in dB, also. Those two graphs are similar enough that it could easily be concluded that the 12AX7 is responding to the parallel impedance of Ck and Rk in the same manner as replacing that impedance with a lone resistor of the same value.
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One time I calculated
I thought I got it, but I didn't :BangHead:
...... :think1:
the 4th read I got :icon_biggrin:
at those frequencies
fwiw when I get really bored I look at this math, then spend 20 minutes hunting up the graph around here that already did the math, I just can't find that one :BangHead:
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mu is not 1600uA/V, transconductance (gm ) is.
Correct. Sorry, that was a minor typo on my part.
1/gm is not the internal cathode resistance of a common-cathode triode gain stage
Oh yes, it is (approximately)! More on this in a moment. :icon_biggrin:
but it is an approximation of the internal cathode resistance for a cathode follower.
It is, because it is exactly the same thing we're talking about! The impedance seen from the outside "looking into" the cathode is exactly the same thing as the output impedance of a cathode follower, except insofar as it's modified a bit by the presence of an external anode resistance (Ra). (More on that in a moment, too.)
The internal cathode resistance for the common-cathode triode gain stage is (Rp + rp ) / (u+1).
Let's think that through for a minute. There are two terms, Rp/(mu+1), and rp/(mu+1). In high-mu triodes like the pair in a 12AX7, the "+1" in (mu+1) is negligible compared to mu itself, and far less than statistical parameter variations, so we can drop it.
So we now have Rp/mu + rp/mu.
And what is the second term, rp/mu?
We know mu = gm x rp. Divide both sides of the equation by rp, and you get (mu/rp) = gm.
Now take the reciprocal of both sides, and you have (rp/mu)=1/gm.
Ta-da! You say pot-eh-to, I say pot-ah-to; you say (rp/u), I say (1/gm). They are the same thing! :icon_biggrin:
I did mention that the external anode resistance Ra has an effect, but didn't include it in the approximate formula. I should have, as it is sometimes comparable to the (1/gm) term.
One time I calculated the parallel impedance of Rk and Ck for a bunch of frequencies and plugged that value into the gain formula in place of Rk . I converted the results to dB using the fully-bypassed gain as the reference point and graphed it. I then took an actual 12AX7 and observed the actual gain at those frequencies and graphed that in dB, also. Those two graphs are similar enough that it could easily be concluded that the 12AX7 is responding to the parallel impedance of Ck and Rk in the same manner as replacing that impedance with a lone resistor of the same value.
You overlooked an error of at least a factor of two. It's easy to overlook a factor of two error on a log-log graph, because of the way a logarithmic axis compresses bigger numbers. On top of that, what tolerance was your capacitor? If it was an electrolytic, it's not unusual to have a -50%/+100% tolerance. That's another huge error.
Try the same experiment again, this time with a 15k cathode resistor going to a negative supply rail (say -15V so the triode still biases up to roughly 1 mA cathode current), and a 1% film cap for the bypass cap, and see what your measurements say. The glaring error in your formula should be clearly visible this time.
This particular error - focusing on the external resistor rather than the combination of external and internal - also plagues many books on transistor electronics. There you frequently see the formula f = 1/(2 pi Re Ce), which is completely wrong; with transistors circuits, the internal emitter resistance is usually far, far smaller than the external resistance, and dominates the frequency response.
-Gnobuddy
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...shelved response...shelving filter...high-pass filter has a shelved response...it is not a shelf filter....A low shelf filter...has a shelved response...
I'm sorry man, you're using a bunch of terms that don't exist in any legitimate electronics textbook, and they make no sense at all to me. "Low shelf filter", for example, makes no sense whatsoever, and has no engineering definition.
But call it by whatever name you want (let's say you want to call it a Pruggle-Flerd filter), the end result is exactly the same: the frequency response caused by a cathode bypass cap is always a flat region at low frequencies, a flat region at high frequencies, and a rising ramp in between.
This is NOT the same as the frequency response of a high-pass filter. The attached image shows the frequency response of an ideal high-pass filter. As you can see, it does NOT flatten out at low frequencies, but falls forever. There is no lower shelf. (A practical implementation of an RC filter will eventually flatten out when it hits the noise floor of your measurement equipment, hopefully at least 90 dB down from the passband region. Very different from a triode with a bypassed cathode cap, which will typically flatten out just a few decibels down from the upper passband region.)
There is more information about the meaning of "high-pass filter" here: https://www.electronics-tutorials.ws/filter/filter_3.html
I hope that helps, and Happy Pruggle-Flerd Filter day to you! :icon_biggrin:
-Gnobuddy
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I'm sorry man, you're using a bunch of terms that don't exist in any legitimate electronics textbook, and they make no sense at all to me. "Low shelf filter", for example, makes no sense whatsoever, and has no engineering definition.
I'm making up terms that don't exist? I was quoting you.
Anybody that has done audio editing on a computer to any extent knows what a low shelf filter is. But you had me doubting myself, so I ran a search on it. Turns out a low shelf filter is exactly what I said. I guess I still have a few memory cells left.
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Let's think that through for a minute. There are two terms, Rp/(mu+1), and rp/(mu+1). In high-mu triodes like the pair in a 12AX7, the "+1" in (mu+1) is negligible compared to mu itself, and far less than statistical parameter variations, so we can drop it.
So we now have Rp/mu + rp/mu.
And what is the second term, rp/mu?
We know mu = gm x rp. Divide both sides of the equation by rp, and you get (mu/rp) = gm.
Now take the reciprocal of both sides, and you have (rp/mu)=1/gm.
Ta-da! You say pot-eh-to, I say pot-ah-to; you say (rp/u), I say (1/gm). They are the same thing!
I did mention that the external anode resistance Ra has an effect, but didn't include it in the approximate formula. I should have, as it is sometimes comparable to the (1/gm) term.
Right out of the gate, the expression involves tomatoes, not potatoes.
You can't just drop Rp out of the equation. Even when the Rp in the equation is replaced by Rp in parallel with the typical 1M grid leak resistor of the following stage, it is still larger than rp. 1/gm is the internal cathode resistance of a cathode follower because Rp is zero and it drops out of the equation.
Isn't the last sentence an admission that you are wrong?
You overlooked an error of at least a factor of two. It's easy to overlook a factor of two error on a log-log graph, because of the way a logarithmic axis compresses bigger numbers. On top of that, what tolerance was your capacitor? If it was an electrolytic, it's not unusual to have a -50%/+100% tolerance. That's another huge error.
Oh no, I didn't.
Listen son, this isn't my first rodeo. I selected a film 680nF cap that tested out real close to 680nF and a 1.5K resistor that was dead-on 1.5K. I have a large collection of 12AX7's on hand, so I selected one that had characteristics that were similar to published values. There is no place to make an error of at least a factor of two when using the formula for the gain of an unbypassed triode. Rk appears in the formula and rk doesn't, nor does the parallel combination of Rk and rk if that is what you are implying. I have it under good authority that rk is already factored into the formula for unbypassed gain.
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Listen son, this isn't my first rodeo.
My daddy's bigger than your daddy.
Have a nice day!
-Gnobuddy
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So how do we calculate the proper bypass cap value?
I say that the formula f = 1/(2 pi Rk Ck) is NOT correct for a cathode bypass cap, and we need a different formula. Arguing about the formula wasn't helping to clarify the issue, or find a solution. So I took a different tack, and used the LTSpice circuit simulator to demonstrate the problem with the formula.
In time honoured mathematical methodology, let's start out by supposing that the formula actually is correct. In that case, as long as we keep the product (Rk x Ck) the same, we should get identical frequency responses, because 1/(2 pi Rk Ck) would be exactly the same.
So, for example, I could use 1.5k and 22uF in one gain stage, then double the resistor to 3k and halve the capacitor to 11uF in a second gain stage. In both cases, the product Rk times Ck is the same (1.5k x 22 uF = 33 milliseconds; 3k x 11 uF = 33 milliseconds; both are the same.) So if our formula is correct, both gain stages should have identical frequency responses.
The attached image shows an LTSpice simulation of two triode gain stages. Both triodes are identical (half a 12AX7), and in the artificial world of the simulator, they are truly identical - there are no normal manufacturing tolerances! The anode resistor and grid bias resistor are also the same for both stages at 100k and 1 megohm respectively.
In the first of the two gains stages, I've used 1.5k and 4.7uF for Rk and Ck. For the second gain stage, I've used ten times the resistance (15k), and one-tenth the capacitance (2.2uF). This guarantees that the product Rk x Ck is identical in both cases, at 7.05 milliseconds. To keep both triodes operating at exactly the same point on the characteristic curves, I've connected a negative supply voltage to the far end of the 15k cathode resistor, and adjusted the voltage so that both triodes show exactly the same current (1.36 mA ) in the simulation.
Both triodes are powered from the same 400V B+ supply. Both anode load resistors are identical (100k). Both anode currents are identical (1.36 mA). This means both anode voltages are also identical (264V in the simulation). Both cathode voltages are identical (+2V in the simulation). Everything is identical in both stages, including the product Rk x Ck. The only difference is that one stage has a ten-times-bigger Rk, and a ten-times-smaller Ck.
So, if our formula is correct, both gain stages will have identical frequency responses.
But look at the result of the simulation. Both stages have identical gain above 3 kHz. But at low frequencies, the frequency responses are not the same, but very different. The 1.5k/4.7uF stage crosses the 14 dB line at 60 Hz. The 15k/0.47uF stage crosses the same 14dB line at 400 Hz! Even though Rk x Ck is identical in both cases, the 15k / 0.47 uF stage is obviously screaming for a bigger capacitor.
Conclusion: The formula f = 1/(2 pi Rk Ck) is incorrect for cathode bypass capacitors. It predicts a much-too-small value of capacitor in circuits where the cathode resistor is large.
Even when the value of Rk is not unusually large, the formula is wrong - it always predicts a better bass response than you will actually get with that capacitor. But the formula is not as badly wrong when Rk is relatively small, as it is when Rk is much bigger.
(An aside: I set the source voltage to 0.1 volt in the simulator, so the graphs are 20 dB lower than you'd see if I had used 1 volt input. The output voltage above 3 kHz is about +15 dB, but the voltage gain at that frequency is 20 dB bigger, at about 35 dB.)
I'm attaching a screenshot of the simulation. If anyone wants the actual LTSpice simulation file (.asc), or the 12AX7 model (.txt), please let me know, and I'll try and upload them here, if the forum permits those file types. That way, you can run the simulation yourself, and verify what it shows.
-Gnobuddy
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Read this:
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:icon_biggrin:
Thanks 2Deaf, my ADHD likes the readers digest version
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Hello again, folks. Haven't gotten much further (financial reasons), but I'm stuck thinking about the PSU bit. Following the advice given, the plan is as follows:
1. Build a 1974x psu, and make it work
2. Build Epi Vjr. stage by stage and make each work
3. Mod - for this I'm thinking I want to add one of the cascode stages from the über-deluxe posted in this thread, a James TS, and convert the second Vjr stage to a cath follower.
Now, I'd like to have a load for each PSU node, so I can take measurements on each before building the respective stage. However, I'm at a loss as to how to figure out what those loads should be... assuming this schematic for the Vjr: http://mercurymagnetics.com/images/pdf/schematics/wiring/E-VJ-schem1a.pdf and going by this statement from PRR A happy SE stage acts-like roughly the OT primary impedance. Use 7K resistor for Champ with 7K OT.
in this thread: https://el34world.com/Forum/index.php?topic=21256.0 - I figure B1 would reasonably be 4.7k @ 25W (that's the closest standard value I've found)? What about B2?
And for B3 I guess I could just do the math for 1mA, considering Merlin's article on PS filtering.
And then for future conundrums, what would be the current draw for a 12ax7 cathode follower? 1mA again? And an 6DJ8/6922/E88CC cascode? This datasheet: http://tdsl.duncanamps.com/dcigna/tubes/sheets/amperex/6922-2g.gif gives a "typical characteric" plate current of 15 mA, should I go with that?
Or is this all overkill? Should I just whack a ~5k 25W on the recto tube and be happy?
Lots of questions here, sorry about that...
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I "test" my PSU at the 1st tap virtually every time, I've got a bunch of 2K N 5K 50W'rs
here's how I do it;
I start with other ppls design :laugh:
grab the tube datasheet
have a "target" 1st tap voltage I'm aiming for.
in this datasheet 250vdc with 1 tube shows average ~~ 48mA @ plate + 5.5mA at G2
so 250vdc/ .04A - ~ 6200 ohms
I gator-clip it to the 1st tap, have my meter clipped across the R and power-up. I quickly grab the Vdc, power down, do math, if I'm happy, I'll powerup and check for things getting warm. If it makes it 5min and no disaster I assume the downstream taps will be ok and start my build
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things got kinda sideways with the cool kids :laugh:
My thinking is, first get the Valve jr/1974 up and running
1. Build a 1974x psu, and make it work
on the recto tube
the schematic you linked has a SS bridge
can you clarify for me or link to your version schematic?
thanks
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> me> A happy SE stage acts-like roughly the OT primary impedance.
That's for transformer loading.
Resistor loading, in ignorance, assume the whole stage acts-like twice the big resistor in the stage (usually plate, but in CF the cathode load).
Using Fender 12AX7-100k-1.5k biasing, it comes to more like 3+ times the plate resistor (so 330k); but "twice" is still a fair guess.
> 6DJ8/6922/E88CC ...a "typical characteric" plate current of 15 mA
That is a TV Tuner tube. It is optimized for 100MHz where everything sucks and noise "snow" is a real issue, and real-high current is the main thing.
We never run these tubes that hot for audio. Do you have a plan? Is it a 47k plate load? Then even if (both halves) the tube were dead-short it could only suck 300V/47k = 6mA. And that won't work. More likely at least half the 300V is dropped in tube(s), so half in the 47k, 3mA. Probably less in cascode.
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Wow, that's two weeks that went "poof"... Anyways,
can you clarify for me or link to your version schematic?
yup, it's the psu from this: https://www.thetubestore.com/lib/thetubestore/schematics/Marshall/Marshall-18-Watt-Schem-Schematic.pdf: (https://www.thetubestore.com/lib/thetubestore/schematics/Marshall/Marshall-18-Watt-Schem-Schematic.pdf:)
Do you have a plan? Is it a 47k plate load?
I snagged the cascode (attached) wholesale from the over-the-top Deluxe Reverb posted previously in this thread: https://sites.google.com/site/stringsandfrets/Home/deluxe-plus (https://sites.google.com/site/stringsandfrets/Home/deluxe-plus). It's bootstrapped by the following CF, upper plate resistor is 100k, lower 68k.
Though that also confuses me, according to Merlin you should be splitting the previous stage's anode resistor into two equal parts
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> two equal parts
68k is 100k, pretty near. There's a bunch of trade-offs which get worse if they are very different, but they do not have to be exactly equal.
Your latest image shows C? at 50V. There's four main parts here: resistor resistor tube tube. On the "roughly equal" theory, each gets ~about~ 1/4 of supply. If supply is 300V, this puts 75V on your 50V part.
I also (again?) suspect there is a high-value grid resistor missing. As drawn, the 820r+1uFd suck-out all the treble current.
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> two equal parts
68k is 100k, pretty near. There's a bunch of trade-offs which get worse if they are very different, but they do not have to be exactly equal.
Your latest image shows C? at 50V. There's four main parts here: resistor resistor tube tube. On the "roughly equal" theory, each gets ~about~ 1/4 of supply. If supply is 300V, this puts 75V on your 50V part.
Oh, on the original schematic, there was no voltage rating on that cap, I'll up it then, thanks.
I also (again?) suspect there is a high-value grid resistor missing. As drawn, the 820r+1uFd suck-out all the treble current.
Bad screenshot/bad idea to put the values inside the symbol... That "R" is actually a "k"
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> Bad screenshot/bad idea
Sorry. My blind.
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Read this:
Read this:
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Good info in here!
The stringsandfrets cascode site is Ulrich Neumann's, not Mr. Aiken's (forgot his first name, but I think he's not Jim Aiken, mon...bad joke). Ulrich redid his website. I couldn't get the earlier link to open but am not sure I see a difference - the one below works for me.
https://sites.google.com/site/stringsandfrets/deluxe-plus
I have been looking at the 1956 (Wireless World?) Hedge amp schematic that uses a cascode LTP with voltage divider bias on the upper triode and wondering how poorly it would work to grid-leak bias the halves of the LTP.
Murray